Dynamic Wireless Power Transfer System for Electric Vehicles to Simplify Ground Facilities, Real-time Power Control and Efficiency Maximization

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Dynamic Wireless Power Transfer System for Electric Vehicles to Simplify Ground Facilities, Real-time Power Control and Efficiency Maximization

Mr. Akshay M Mrs.Veenarani A V

Dept. of Electrical and Electronics Engg. Dept. of Electrical and Electronics Engg Srinivas Institute of Technology, Mangaluru Srinivas Institute of Technology, Mangaluru

Abstract:- Wireless Power Transfer (WPT) is the process where electrical energy is transferred from a power source to an electrical load across an air gap using induction coils. These coils produce an electromagnetic field which sends energy from a charging base station (transmitter) to a coil on a portable device (receiver) with complete galvanic isolation. The receiver coil takes power from electromagnetic field and converts it into electrical power.

This focuses on dynamic wireless power transfer for electric vehicles and proposes a vehicle-side control method for real- time power control and efficiency maximization. The proposed control strategy and controller design are presented based on a real-time estimation of the mutual inductance between a transmitter and a receiver. Simulations and experiments demonstrate that the proposed method can achieve the maximum efficiency and the desired power simultaneously.

  1. INTRODUCTION

    The transfer of energy from a source to a receiver has traditionally necessitated the use of a physical connection. Indeed, electrical grids and the power outlets span nearly the entire globe and deliver power to billions of people worldwide. Recently, there has been much interest into the area of wireless power transfer (WPT) that is the transmission of power without the need for physical connection. Research into WPT, however is nothing new as experiments in the field took place as far back as Nikola Tesla in the early 20th century.

    Wireless power transfer (WPT) is one of the hottest research topics in transportation applications. In particular, a dynamic wireless power transfer (DWPT) system for electric vehicles (EVs) has gathered attention to extend the cruising distance of EVs and to reduce the size of the energy storage system. Its ground facilities are mainly composed of power source, high-frequency inverters, transmitters, and so on. As they are applied to rugged roadways over long distances, power control and efficiency maximization of wireless charging are desirable to be achieved on the vehicle side without signal communication. Although previous research has proposed simultaneous control methods of power control and efficiency maximization on the vehicle side, they have not been

    applied to the DWPT system. For maximizing the transmitting efficiency in the DWPT system, the mutual inductance between a transmitter and a receiver has to be estimated from the vehicle side. In this paper, an estimation method considering the vehicle-side control is proposed and applied to the simultaneous power and efficiency control. The effectiveness of the proposed method is verified by simulations and experiments.

  2. WIRELESS POWER TRANSFER VIA MAGNETIC RESONANCE COUPLING

    1. Circuit analysis:

      The transmitter and the receiver coils are shown in Fig. 2.1. They are compensated by resonance capacitors for WPT via magnetic resonance coupling, which can achieve a highly efficient mid-range transmission and robustness to misalignment. In this paper, a series-series (SS) compensated circuit topology is used and its circuit diagram is shown in Fig. 2.2. The transmitter and the receiver are characterized by the inductances L1, L2, the series-resonance capacitances C1, C2, and the internal resistances R1, R2, respectively.Lmis the mutual inductance between the transmitter and the receiver.

      These parameters are expressed in Table 2.1. If power source angular frequency is designed as follows:

      F ig 2.1: Coil

      Table 2.1 : Parameters of the coil

      R = R2 (0Lm)2 + R2} — (2.4)

      Transmitter

      Receiver

      Resistance R1, R2

      1.95

      1.6

      Inductance L1, L2

      417.1µH

      210µH

      Capacitance C1, C2

      6030pF

      12110pF

      Resonant frequency f1, f2

      100.4kHz

      99.7kHz

      Coil gap

      100mm

      Mutual Inductance Lm

      36.3µH (no misalignment)

      Coupling coefficient k

      0.122 (no misalignment)

      Transmitter

      Receiver

      Resistance R1, R2

      1.95

      1.6

      Inductance L1, L2

      417.1µH

      210µH

      Capacitance C1, C2

      6030pF

      12110pF

      Resonant frequency f1, f2

      100.4kHz

      99.7kHz

      Coil gap

      100mm

      Mutual Inductance Lm

      36.3µH (no misalignment)

      Coupling coefficient k

      0.122 (no misalignment)

      Lmax { R1

      Fig. 2.3: Equivalent circuit of WPT system

      Then, the transmitting power P is determined byRmax. As a result, the desired power cannot be achieved only using RL optimization when the transmitting efficiency is maximized.

    2. System configuration:

      In order to achieve the maximum efficiency and the desired power simultaneously, the vehicle is equipped with two power converters, i.e. Half Active Rectifier (HAR) and the DC-DC converter. The circuit diagram of the DWPT system is shown in Fig. 2.4. The ground facility consists of voltage source VS and an inverter, which generates a square wave voltage with resonance angular frequency0. The transmitting power P is rectified by the HAR and the charging power PL is controlled by the DC-DC converter. These control strategies and controller design methods are

      described below.

      Fig 2.5: Circuit diagram of the DWPT system.

      = 1

      0 L1C1

      Fig. 2.4: RL v/s and P

      = 1 — (2.1)

      L2C2

  3. Efficiency Maximization by Half Active Rectifier

    1. DC link voltage control:

      The HAR regulates the DC link voltageVdcfor efficiency maximization.VdcControl is achieved using two operation modes of HAR, which are shown in Fig. 3.1. When the lower arm MOSFETs are off-state, HAR is operated as the rectification mode. If the MOSFETs are turned on, HAR becomes the short mode and the receiver is shorted. Assuming the transmitting power P is larger than the load power PL,Vdcis increased during the rectification mode. On the other hand,Vdcis decreased during the short mode because P is cut-off and PL is supplied by the DC link

      The transmitting efficiency and the transmitting power P can be obtained as follows:

      capacitor. Therefore, the waveform ofVdccan be depicted in Fig.3.2.

      (0Lm)2 RL

      =

      R2 + RL{R1R2 + R1RL + (0Lm)2}

      — (2.2)

      2

      2

      P= (0Lm) RL

      {R1R2 + R1RL + (0Lm)2}2

      V12 — (2.3)

      Where V1 is the RMS value of the primary voltage and RL is the load resistance. When VI equals to 100V, and P are expressed in Fig 2.3. In order to maximize the transmitting efficiency , RL has to be given as follows:

      eq. (3.4), output y[i] and regressor[i] are determined as follows:

      [i] = V1 + V12 4R1I2[i](V2[i] + R2I2[i])— (3.5)

      [i] = 2I2[i]0 — (3.6)

      RLS filter is used to estimate Lm statistically by updating

      Lm[i], y[i] and [i] with following equations.

      Lm[i] = Lm[i 1] + [i]P[i1]

      + [i]2P[i1] [i] [i] = y[i] [i]Lm[i 1]

      2 2

      2 2

      P[i] = 1 {P[i 1] [i] P[i1] } — (3.7)

      +[i]2P[i1]

      Fig 3.1: Operation modes of Half Active Rectifier. Fig 3.2: Waveform of the DC link voltage

      If the upper boundVhigh and the lower boundVloware defined as follows:

      Vhigh = Vdc + V — (3.1)

      Vlow = Vdc V — (3.2)

      WhereVdcthe reference value of the DC link voltage and V is the hysteresis band,Vdc is kept within the desired range.

    2. Efficiency Maximization:

      In order to maximize the transmitting efficiency, the load resistance RL, which is expressed in Fig. 2.2 has to satisfy eq. (2.4) during the rectification mode. IfVdcis given as follows:

      Where is forgetting factor. The initial values are given as Lm[0] = 0 and P[0] = 1.In order to use the effective values for the estimation, the RLS filter updates Lm[i], y[i] and [i] only during the rectification mode of the HAR. If the HAR is operated as the short mode,Idcequals to 0 andthe estimated value becomes incorrect. Therefore, the RLS filter has to be improved according to theoperation mode of the HAR.

  4. POWER CONTROL BY THE DC-DC CONVERTER

    1. Modeling of the DC-DC converter:

      The DC-DC converter controls the load current iLfor battery charging. Assuming the DC link voltage Vdcis controlled to Vdcmax by the HAR, the circuit diagram of the DC-DC converter can be expressed in Fig. 4.1 (a). E is the

      V = R2 0Lm V

      — (3.3)

      battery voltage, L is the inductance of the reactor coil and r

      dcmax

      R1

      R1R2+(0Lm)2

      S

      +R1R2

      is the internal resistance of the reactor coil and the battery. In this paper, the DC-DC converter model is obtained by the state space averaging method. Assuming the DC-DC

      RL is equated toRLmaxand the transmitting efficiency can

      be maximized. On the other hand, during the short mode, the transmitting power P is drastically decreased because RL is minimized. As a result, losses during the short mode are assumed to be negligible to losses during the

      rectification mode in this paper. Therefore,Vdc is determined only byVdcmax.

      3.3 Mutual inductance estimation:

      For tracking the maximum efficiency in the DWPT system, the mutual inductance Lm has to be estimated to obtainVdcmax.only using the vehicle-side information. From the circuit equations of the DWPT system, the estimation equation of Lm can be given as follows.

      V1+V124R1I2(V2+R2I2)

      converter is operated in the continuous conduction mode, its circuit diagram in each switching modes is expressed in Fig. 4.1 (b) and (c).

      Figure 4.1: Circuit diagram of the DC-DC converter.

      Lm =

      2I20

      — (3.4)

      Although eq. (3.4) has two solutions, the solution with a positive sign is used in this paper. Assuming the RMS

      From the circuit equation, the state equation of Fig.4.1 (b) is given as follows:

      primary voltage V1 is constant and given to simplify

      d i (t) = r i (t) + 1 E + 1 V

      —(4.1)

      ground facilities, Lm can be estimated from the vehicle

      dt L

      L L L

      L dcmax

      side. The RMS secondary voltage V2and the RMS secondary current I2are calculated from the DC link

      Also, the state equation of Fig.4.1 (c) is described as follows:

      d i (t) = r i (t) + 1 E — (4.2)

      voltageVdcand the rectified DC currentIdcusing Fourier

      dt L

      L L L

      series expansions.

      In order to reduce the estimation error due to the sensor noise, recursive least square (RLS) filter is applied. From

      When d(t) is defined as the duty cycle of the upper arm MOSFET S1, the state space model of the DC-DC converter is obtained as follows:

      d i (t) = r i (t) + 1 E + Vdcmax dt — (4.3)

      The feedback controller is designed by the pole placement

      dt L

      L L L L

      method. As Pi(s)is the first-order system, we apply a PI

      In order to apply the linear control theory to the controller

      controllerCPI(s), which is expressed as follows:

      design, this model is linearized around an equilibrium

      sKP+KI

      C (s) =

      C (s) =

      PI s

      — (4.10)

      point. When IL and D are defined as the equilibrium point,iL(t) and d(t) are expressed as follows:

      iL(t) = IL+iL(t) ` — (4.4)

      If closed loop poles are given by a multiple rootd, the gains are obtained as follows:

      d(t) = D+d(t) — (4.5)

      wherei (t)and d(t) are the microscopic fluctuations

      K = 2Ldr P Vdcmax

      — (4.11)

      around

      L

      the equilibrium point.

      KI =

      Ld2

      — (4.12)

      By substituting eq. (4.4) and eq. (4.5) in eq. (4.3), the linearized DC-DC converter model is given as follows:

      d i (t) = r i (t) + Vdcmax dt — (4.6)

      Vdcmax

      Here,Vdcmaxand the gains are calculated by assuming the nominal value of Lmis 30h

      dt L L L L

      Therefore, the transfer function from d(s) to iL(s) is obtained as follows:

  5. SIMULATION AND EXPERIMENT

    5.1 Experimental setup:

    Pi(s) = iL(s) = Vdcmax

    — (4.7)

    The experimental setup is shown in Fig. 5.1. The system

    d(s)

    Ls + r

    configuration is the same as Table 5.1. The receiver is

    4.2 Controller design:

    Fig.4.2 shows the block diagram of the proposed control. The feed-forward controller is the same as the equilibrium point calculation, which is given by the constraint of the DC-DC converter. FromVdcmax,which is calculated from

    Lm, and the reference value of the load currentiL, the equilibrium point isobtained as follows:

    IL = iL — (4.8)

    driven by the motor to simulate motion of the vehicle. The inverter is operated only while the receiver is above the transmitter to prevent huge power losses. Simulation and experimental conditions are expressed in Table 5.1. The forgetting factor of the RLS filter was set to 0.95 and the estimated mutual inductance Lm was updated only during the rectification mode of the HAR. The reference value of the DC link voltageVdcmaxwas calculated from Lm and the

    reference value of the load currenti was set to 1.0 A.

    D = E+rIL

    Vdcmax

    L

    — (4.9)

    Fig. 4.2: Block diagram of the proposed control.

    Parameter

    Value

    Parameter

    Value

    Power Source Voltage VS

    18V

    Battery voltage E

    6V

    Operating Frequency f0

    100kHz

    Reactor resistance r

    0.3

    Hysteresis band V

    0.5V

    Reactor Inductance L

    1000µH

    Carrier frequency fc

    20kHz

    DC Link capacitance Cdc

    200pF

    Parameter

    Value

    Parameter

    Value

    Power Source Voltage VS

    18V

    Battery voltage E

    6V

    Operating Frequency f0

    100kHz

    Reactor resistance r

    0.3

    Hysteresis band V

    0.5V

    Reactor Inductance L

    1000µH

    Carrier frequency fc

    20kHz

    DC Link capacitance Cdc

    200pF

    Fig. 5.1: Experimental setup.

    Fig. 5.3: Simulation results with the proposed control.

    Table 5.1: Simulation and experimental conditions.

      1. Simulation:

        In the simulations, the chnge in Lm was simulated by a sine wave. Its minimum and maximum values were set to 20H and 40H.

        Fig. 5.2: Simulation results without the proposed control

        Fig.5.2 shows the simulation results without the proposed control. In this simulation, the HAR was operated as only the rectification mode and the duty cycle d of the DC-DC converter was equated to 0.95. From Fig.5.2 (b), Lm is closely matched with the actual Lm. However, the transmitting efficiency is decreased from the maximum efficiency because the DC link voltageVdccannot be regulated toVdcmax. Furthermore, the load currentiLcannot

        be controlled toiL. Fig. 5.3 shows the simulation results with the proposed control. The closed loop poles of the

        proposed control were placed at -2000 rad/s. Although Lm was estimated only during the rectification mode of the HAR, Lm accords with the actual Lm as shown in Fig.5.3 (b). From Fig.5.3 (c) and (d),Vdcis regulated aroundVdcmaxand is maximized during the rectification mode. In addition, Fig.5.3 (e) indicates that the load current control can be achieved.

      2. Experiment:

    In the experiments, the receiver was driven at 10 km/h and Lm was compared to the actual Lm, which was measured by an LCR meter (NF Corp., ZM2371). The DC to DC efficiencydcincludes not only the transmitting efficiency but also the converter efficiency because it was measured by the DC voltages and currents on each side. Therefore, improvement of system efficiency is verified in the experiments. Fig.5.4 shows the experimental results

    without the proposed control. The HAR and the DC-DC converter were operated at the same condition as the simulation without control. From Fig.5.4 (b), Lm and the actual Lm are closely matched. Although Lm has a short- time delay,Vdcmax is near by the actualVdcmax as shown Fig.5.4(c). However, the transmitting efficiency cannot be maximized becauseVdcis not regulated toVdcmax. Furthermore, Fig. 5.4 (e) indicates thatiLcannot be controlled unless the proposed control is applied. In the case of with control, the DC-DC converter started power control whenVdcreachedVdcmax. The closed loop poles of

    the proposed control were placed at -1000 rad/s. Fig. 5.5 shows the experimental results with the proposed control. Although the error of Lm is larger than without control,Vdccan be controlled aroundVdcmax as shown in Fig. 5.5 (c). From Fig. 5.5 (d),dcduring the rectification mode of the HAR is increased compared to without control. In addition, Fig. 5.5 (e) shows thatiLcan be controlled toiL. If the closed loop poles of the proposed control are optimized, it is possible to suppress the current ripple due

    to the fluctuation ofVdc.

    Fig. 5.4: Experimental results without the proposed control

    Figure 5.5: Experimental results with the proposed control

  6. CONCLUSION

This proposed a simultaneous control method of real-time power control and efficiency maximization based on improved mutual inductance estimation from the vehicle side. Its control strategy and controller design methods were presented. The effectiveness of the proposed method was verified by the simulations and the experiments. Future works are to propose an efficiency maximization method considering losses during the short mode of HAR and to design an optimal controller for the proposed control. Furthermore, the proposed method is implemented to an actual DWPT system using an EV.

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